Multiply-connected power processing

ABSTRACT

Various power processing systems are described that employ a multiply-connected velocity inhibiting circuit. At least one active circuit can be employed to synthesize at least one passive lumped element in the multiply-connected velocity inhibiting circuit.

CROSS REFERENCE TO RELATED APPLICATION

This application is a continuation of, and claims priority to,co-pending U.S. patent application entitled “MULTIPLY-CONNECTED POWERPROCESSING” filed under Ser. No. 13/664,978 on Oct. 31, 2012, which is acontinuation of, and claims priority to co-pending U.S. patentapplication entitled “MULTIPLY-CONNECTED POWER PROCESSING” filed underSer. No. 12/437,041 on May 7, 2009, which claims priority to U.S.Provisional Patent Application entitled “MULTIPLY-CONNECTED POWERPROCESSING” filed under Ser. No. 61/051,388 on May 8, 2008, all three ofwhich are incorporated herein by reference in their entirety.

BACKGROUND

Power distributions systems such as the North American power grid can besubject to many anomalies that degrade the quality of the powergenerated. For example, such anomalies may include transients,harmonics, black outs, brown outs, voltage surges, voltage sags, andother anomalies may occur that can adversely effect sensitive electricalloads. For example, many merchants that sell products on the Internetmake use of large server banks that not only need clean power, but needto be protected from outages that might adversely affect the ongoingsales transactions.

BRIEF DESCRIPTION OF THE DRAWINGS

Many embodiments of the invention can be better understood withreference to the following drawings. The components in the drawings arenot necessarily to scale, emphasis instead being placed upon clearlyillustrating the principles of the various embodiments of the presentinvention. Moreover, in the drawings, like reference numerals designatecorresponding parts throughout the several views.

FIG. 1 is a drawing of a power multiplier according to the prior art;

FIG. 2 is a drawing of a directional coupler of the power multiplier ofFIG. 1;

FIG. 3 is a drawing of an impractical power multiplier with respect to ageographical map illustrating a problem of practicing powermultiplication using a power multiplier illustrated in FIG. 1 at powerfrequencies of relatively small wavelengths;

FIG. 4A is a block diagram of power transmission line from a powergenerator to an electrical load;

FIG. 4B is a schematic of an equivalent impedance per length oftransmission line of FIG. 4A;

FIG. 5 is a drawing of alternative transmission lines that might beemployed as the power transmission line of FIG. 4A and that have anequivalent impedance that can be modeled by the schematic of FIG. 4B;

FIG. 6A is a schematic of a T-network employed in a power multiplieraccording to one embodiment;

FIG. 6B is a schematic of a 7-network employed in a power multiplieraccording to one embodiment;

FIG. 7A is a schematic of an embodiment of the T-network of FIG. 6A;

FIG. 7B is a schematic of an embodiment of the π-network of FIG. 6B;

FIG. 8 is a schematic of a power multiplying network according to oneembodiment;

FIG. 9 is a schematic of a phase shifter employed in the powermultiplier of FIG. 8 according to one embodiment;

FIG. 10 is a schematic of a directional coupler employed in the powermultiplier of FIG. 8 according to one embodiment;

FIG. 11 is schematic of a power multiplying network according to oneembodiment;

FIG. 12 is a schematic of a power multiplying network according to oneembodiment;

FIG. 13 is a schematic of one example of a multiply-connected powerprocessor according to one embodiment;

FIGS. 14A and 14B depict examples of a gyrator circuit employed as anactive circuit in the multiply-connected power processor of FIG. 13according to one embodiment;

FIG. 15 is a schematic of one example of a source coupling employed inthe multiply-connected power processor of FIG. 13 according to oneembodiment;

FIG. 16 is a schematic of one example of a load coupler employed in themultiply-connected power processor of FIG. 13 according to oneembodiment;

FIG. 17 is a schematic of one example of a phase shifter employed in themultiply-connected power processor of FIG. 13 according to oneembodiment;

FIG. 18 is a schematic of cascaded second order low pass filter cells ofthe power multiplying network, for example, of FIG. 8 according tovarious embodiments;

FIG. 19 is a schematic of another example of a multiply-connected powerprocessor according to another embodiment;

FIG. 20 is a schematic of an active circuit employed in themultiply-connected power processor of FIG. 19 according to oneembodiment;

FIG. 21 is a schematic of one example of a source coupling employed inthe multiply-connected power processor of FIG. 19 according to oneembodiment;

FIG. 22 is a schematic of one example of a load coupler employed in themultiply-connected power processor of FIG. 19 according to oneembodiment; and

FIG. 23 is a schematic of another example of a multiply-connected powerprocessor employed for power or frequency conversion according tovarious embodiments.

DETAILED DESCRIPTION

With reference to FIG. 1, shown is a power multiplier 100 according tothe prior art. The power multiplier 100 includes a power multiplyingwaveguide 103 and a launching waveguide 106. Both the power multiplyingwaveguide 103 and the launching waveguide 106 are conventionaltransmission lines such as hollow pipes, coaxial cables, parallel wiretransmission lines. The launching waveguide 106 is coupled to the powermultiplying waveguide 103 using a directional coupler 109. Anelectromagnetic signal generator 113 is coupled to the launchingwaveguide 106 and generates an exciting traveling wave 116 that islaunched into the launching waveguide 106. The directional coupler 109includes two slits 119 that are spaced apart by distance D. The distanceD is approximately equal to ¼ of wavelength of the exciting travelingwave 116. Thus, the electromagnetic signal generator 113 generates theexciting traveling wave 116 at a predefined frequency having awavelength λ_(w) that is approximately four times the electricaldistance D/λ_(w). The launching waveguide 106 terminates in a matchedload 123. The total length of the power multiplying waveguide 103 is aninteger multiple of the wavelength λ_(w) of the exciting traveling wave116. In the case that the power multiplying waveguide 103 is a closedcircle or closed ring as shown, the total length of the powermultiplying waveguide is equal to its circumference.

To operate the power multiplier 100, the electromagnetic signalgenerator 113 generates the exciting traveling wave 116 that is launchedin the launching waveguide 106. When the exciting traveling wave 116reaches the directional coupler 109, a portion of the exciting travelingwave 116 is coupled into the power multiplying waveguide 103, therebycreating a traveling wave 126 that propagates along the powermultiplying waveguide 103. The directional coupler 109 couples theportion of the exciting traveling wave 116 into the power multiplyingwaveguide 103 in such a manner that the traveling wave 126 travels in asingle direction around the power multiplying waveguide 103.Specifically, since the distance D between the slits 119 isapproximately equal to ¼ of the wavelength λ_(w) of the excitingtraveling wave 116, all energy coupled into the power multiplyingwaveguide 103 propagates in a single direction as will be furtherdescribed with reference to later figures.

In addition, since the length of the power multiplying waveguide 103 isan integer multiple of the wavelength λ_(w) of the exciting travelingwave 116, the traveling wave 126 is spatially synchronized with theexciting traveling wave 116. Under these conditions, the portion of theexciting traveling wave 116 that is continually coupled into the powermultiplying waveguide 103 reinforces or is added to the traveling wave126. Consequently, the power of the traveling wave 126 may become quitelarge in magnitude. That is to say, the Poynting's vector power flow,½Re{E×H*} is pumped up within the power multiplying waveguide, which isa linear, passive, distributed energy storage structure. The averageenergy of the traveling wave 126 is “distributed” in that it is evenlydistributed throughout the entire length of the power multiplyingwaveguide 103.

Once begun, the buildup of the power of the traveling wave 126 withinthe power multiplying waveguide 103 will continue until the lossesaround the power multiplying waveguide 103 plus the loss in the matchedload 123 that terminates the launching waveguide 106 is equal to thepower generated by the electromagnetic signal generator 113. The powermagnification M and optimum coupling C_(Opt) may be calculated asfollows:

${M = \frac{1}{( {1 - A^{2}} )}},{and}$ C_(Opt) = 1 − A²,

where A is the field propagation decay for a single traversal of thepower multiplying waveguide 103. The quantity of C_(Opt) is that valueof coupling for which the magnification is maximized.

The directional coupler has the property that energy leaking from thepower multiplying waveguide 103 back into the launching waveguide 106 isreduced in magnitude. Also, energy leaking back into the launchingwaveguide 106 propagates only in a single direction towards the matchedload 123 and, since such energy is of the correct phase, it cancels outthe power propagating from the electromagnetic signal generator 113 tothe matched load 123. Consequently, when the exciting traveling wave 116and the traveling wave 126 are in phase, the matched load 123 dissipateslittle or no power. Convenient nomograms for the engineering design oflossy power multipliers operating at ultra-high frequencies aredescribed in Tomiyasu, K., “Attenuation in a Resonant Ring Circuit,”IEEE Transactions on Microwave Theory and Techniques, Vol. MTT-8, 1960,pp. 253-254.

Referring next to FIG. 2, shown is a drawing of a portion of the powermultiplying waveguide 103 and a portion of the launching waveguide 106.Also shown is the directional coupler 109. The drawing of FIG. 2 isprovided to further explain the function of the directional coupler 109.To explain the operation of the directional coupler 109, the excitingtraveling wave 116 is launched into the launching waveguide 106 andapproaches the first slit 119 a. A portion of the exciting travelingwave 116 enters the power multiplying waveguide 103 through the firstslit 119 a propagates in both directions within the power multiplyingwaveguide 103 as wave portion W₁ and wave portion W₂. The portion of theexciting traveling wave 116 that does not pass through the first slit119 a proceeds along the launching waveguide 106 until it reaches thesecond slit 119 b. At this point, a second portion of the excitingtraveling wave 116 enters the power multiplying waveguide 103 throughthe second slit 109 b and propagates in both directions in the powermultiplying waveguide 103 as wave portion W₃ and wave portion W₄. If thedistance D between the slits is equal to ¼ of the wavelength λ_(w) ofthe exciting traveling wave 116 as shown, then the wave portion W₃cancels out the wave portion W₁. Also, the wave portion W₂ reinforcesthe wave portion W₄, thereby resulting in the traveling wave 126. As aconsequence of the cancellation of wave portions W₁ and W₃, and thereinforcement of wave portions W₂ and W₄, the traveling wave 126proceeds in a single direction around the power multiplying waveguide103. Given that the exciting traveling wave 116 and the traveling wave126 are in phase or are spatially synchronized, the portion of theexciting traveling wave 116 that is coupled into the power multiplyingwaveguide 103 is continually added to the traveling wave 126, therebymultiplying the power of the traveling wave 126. The power of thetraveling wave 126 is real power. This is to say that there is noreactive component.

Referring next to FIG. 3, shown is a drawing of a map of the UnitedStates 133 that illustrates the problem that prevents the operation ofpower multipliers 100 at low frequencies such as power frequencies.Assume, for example, that the frequency of operation is 60 Hertz whichrepresents the frequency of the power generation system of the UnitedStates. Assuming that the speed of light is approximately 300,000km/sec, at 60 Hertz, the wavelength of both the exciting traveling wave116 and the traveling wave 126 is calculated as:

$\lambda_{w} = {\frac{c}{f} \approx \frac{300,000\mspace{14mu} {km}\text{/}\sec}{60\mspace{14mu} {Hz}} \approx {5000\mspace{14mu} {{km}.}}}$

Thus, the length or circumference of a hypothetical power multiplyingwaveguide 100 a would have to be approximately 5000 Kilometers.Consequently, a corresponding hypothetical transmission line 101employed in the power multiplying waveguide 100 a would be approximately5000 Kilometers in length. Obviously, due to the size involved, thecreation of such a power multiplying waveguide 100 a is not physicallypractical and is cost prohibitive.

Turning then to FIG. 4A, we turn our attention to a discussion of powertransmission lines. In FIG. 4A, a power generator 153 is electricallycoupled to an electrical load 156 by a power transmission line 159. Sucha transmission line 159 may be traditionally employed, for example, todistribute power to homes and businesses as can be appreciated by thosewith ordinary skill in the art.

Referring next to FIG. 4B, shown is an equivalent circuit 163 thatillustrates the equivalent impedance per unit length of the transmissionline 159 (FIG. 4A). Specifically, each unit length of the transmissionline 159 includes series inductance L_(T) and series resistance R_(T).Also, between the conductors of the transmission line 159 are a shuntcapacitance C_(T) and a shunt conductance G_(T). Accordingly, theequivalent impedance per unit length of the transmission line 159 may beexpressed in terms of a series inductance L_(T), a series resistanceR_(T), a shunt capacitance C_(T), and a shunt resistance R_(T).

The equivalent circuit 163 reflects that fact that transmission lines159 direct the propagation of field energy. The field energy propagatingalong a transmission line 159 is stored in the magnetic fields andelectric fields associated with the structure of the transmission line159 itself. On a mode-by-mode basis, one can equate the magnetic fieldenergy stored in a transmission line 159 to the magnetic field energystored in an equivalent distributed inductance. Also, the energy storedin the electric fields of the line can be equated to the energy storedin an equivalent distributed capacitance. Field power losses per unitlength of the transmission line 159 can be equated to the equivalentseries resistive and shunt conductive losses per unit length.

Turning then to FIG. 5, shown are various embodiments of thetransmission line 159 (FIG. 4A) for which the equivalent impedance maybe expressed using the equivalent circuit 163 (FIG. 4B) discussed above.For example, transmission line 159 may comprise, for example, a paralleltransmission line 159 a that includes parallel conductors 166.Alternatively, the transmission line 159 may comprise a coaxialtransmission line 159 b that includes an inner conductor 169 and anouter conductor 173. In yet another alternative, the transmission line159 may comprise an electrical structure 159 c that includes a conductor176 of a predefined geometry situated with respect to a ground plane179. Alternatively, the conductor 176 may be situated with respect to asecond such conductor rather than the ground plane 179. The predefinedgeometry of the conductor 176 may be, for example, a helix or othergeometry. In still another alternative, the transmission line 159 maycomprise an electrical structure 159 d that comprises a single conductor181 in the form a helix or other appropriate shape. In addition thetransmission line 159 may comprise other types of transmission lines andelectrical structures such as, for example, strip lines, fiber opticcables, and so on as can be appreciated by those with ordinary skill inthe art.

Assuming that were actually possible to create the power multiplier 100a at power frequencies such as 60 Hertz, for example, such a powermultiplier 100 a would involve the use of transmission wire in one ofthe configurations described above. In this respect, the impedance ofsuch a transmission wire can be calculated and the equivalent impedancein terms of the series inductance L_(T) (FIG. 4B), the series resistanceR_(T) (FIG. 4B), the shunt capacitance C_(T) (FIG. 4B), and the shuntconductance G_(T) (FIG. 4B) can be determined.

With reference to FIGS. 6A and 6B, shown are a T-network 183 and aπ-network 186 that may be employed according to the various embodimentsof the present invention. In this respect, the T-Network 183 includesseries impedance Z₁ and series impedance Z₂. The T-Network 183 alsoincludes parallel impedance Z₃. The characteristic impedance Z₀ of asymmetrical T-network 183 shown may be calculated as follows:

Z ₀=√{square root over (Z ₁(Z ₁+2Z ₃))}.

The π-network 186 includes parallel impedances Z_(A) and Z_(B). Theπ-network 186 also includes series or middle impedance Z_(C). Thecharacteristic impedance Z₀ of a symmetrical π-network 186 may becalculated as follows:

$Z_{0} = {Z_{A}{\sqrt{\frac{Z_{C}}{( {Z_{C} + {2Z_{A}}} )}}.}}$

For further discussion of both the T-network 183 and/or the π-network186, reference is made to Terman, F. E., Radio Engineering Handbook,McGraw-Hill, 1943, pp. 172-178, 191-215, which is incorporated herein byreference in its entirety. The T-network 183 and/or the π-network 186may be employed, for example, in the construction of a power multiplieraccording to various embodiments of the present invention as will bediscussed. In particular, the impedance represented by the T-network 183and/or the π-network 186 are forms of the equivalent circuit 163 (FIG.4B).

Referring next to FIGS. 7A and 7B, shown are an example schematic ofboth a T-network 183 a and a π-network 186 a that may be employed invarious embodiments of the present invention. In this respect, theT-network 183 a includes series inductance L that is shown as twoseparate series inductances L/2. In addition, the T-network 183 a alsoincludes a shunt capacitance C. The T-network 183 a includes a seriesloss resistances R and a shunt conductance G that are inherent in theconductors making up the inductances L/2, the capacitance C, and theelectrical wire connecting such components.

The π-network 186 a includes a series inductance L and shuntcapacitances C/2. For multiple π-networks 186 a that are coupledtogether in series, adjacent shunt capacitances C/2 may be addedtogether to become capacitance C. The 7-networks 186 a also includes aseries resistance R and a shunt conductance G that are inherent in theconductors making up the inductance L, the capacitances C/2, and theelectrical wire connecting such components. The T-network 183 a andπ-network 186 a illustrate more particular embodiments of the T-networks183 or π-networks 186.

Turning then, to FIG. 8, shown is an example of a power multiplier 200according to an embodiment of the present invention. The powermultiplier 200 includes a power multiplying network 203 and a launchingnetwork 206. The launching network 206 also includes a directionalcoupler 209 that couples the launching network 206 to the powermultiplying network 203. A power source 213 is coupled to the launchingnetwork 206. Also, the launching network 206 is terminated in a matchingload R_(L).

In one embodiment, the power multiplying network 203 is amultiply-connected, velocity inhibiting circuit constructed from anumber of lumped-elements 216. As contemplated herein, the term“network” is defined as an interconnected structure of electricalelements. The terms “multiply-connected” is a mathematical termdescribing the existence of a closed path in a resonator, waveguide, orother electrical structure that cannot be reduced to a point withoutpart of the closed path passing through regions that are external to thegeometrical boundaries of the resonator, waveguide, or other electricalpathway. The power multiplying network 203 is “velocity inhibiting” asthe electrical structure of the power multiplying network 203 results ina reduced velocity of propagation of an electromagnetic wave through thepower multiplying network 203 relative to the speed of anelectromagnetic wave through free space, which is the speed of light.

In addition, the term “lumped” is defined herein as effectivelyconcentrated at a single location. Thus, the terms “lumped-elements”refer to discrete, two-terminal, concentrated electrical elements suchas capacitance, inductances, resistance, and/or conductance. Thus, thelumped-elements as described herein may comprise discrete inductors,capacitors, or resistors. In addition, as contemplated herein,lumped-elements may also comprise diodes, transistors, and othersemi-conductors that may be described, for example, as nonlinearresistors or conductors that have resistance or conductance that iscontrolled by the polarity of applied voltages or currents, etc. Inaddition, lumped-elements may also comprise inherent capacitances,inductances, resistances, or conductances of various electricalstructures such as helices, parallel plates, or other structure as willbe discussed. Similar to the power multiplying network 203, thedirectional coupler 209 is also constructed using lumped-elements.

The power multiplying network 203 is a velocity inhibiting circuit thatresults in a slower velocity of propagation of an electrical disturbancesuch as a traveling wave. In this respect, the power multiplying network203 has an electrical length that is equal to an integer multiple of thewavelength of the operating frequency of the power source 213. Due tothe velocity inhibited nature of the power multiplying network 203, itssize is quite compact in comparison with the wavelength of the operatingfrequency of the power source 213. In addition, the directional coupler209 causes a phase shift that is equal to one quarter of the wavelengthof an exciting traveling wave generated by the power source 213 at theoperating frequency as will be discussed.

In one embodiment, the power multiplying network 203 is constructed fromlumped-elements 216 such as, for example, the inductances L andcapacitances C as shown in FIG. 8. In one embodiment, the inductances Lmay be actual inductors and the capacitances C may be actual capacitorsthat are either commercially available or may be constructed as needed.For example, the power multiplying network 203 may be characterized as aring of interconnected T-networks 183 a (FIG. 7A) or π-networks 186 a(FIG. 7B), although the interconnected T-networks 183 a (FIG. 7A) orπ-networks 186 a (FIG. 7B) may be arranged in a multiply-connectedstructure other than a ring. Each of the T-networks 183 a or π-networks186 a may be considered a “section” of the power multiplying network203. In this respect, assuming that the power multiplying network 203comprises a number of T-networks 183 a, then each inductance L may bedivided into two series inductances L/2 that make up the seriesinductances L/2 as described in the T-network 183 a (FIG. 7A).Similarly, assuming that the power multiplying network 203 comprises anumber of π-networks 186 a, each capacitance C may be also be viewed asa pair of shunt capacitances C/2, each such shunt capacitance C/2 makingup one of the shunt capacitances C/2 of the π-network 186 a (FIG. 7B).Whether T-networks 183 a or π-networks 186 a are employed to create thesections of the power multiplying network 203, each of the networks 183a or 186 a results in a predefined phase shift φ_(s).

Assuming that either T-networks 183 a or π-networks 186 a are to beemployed to construct the power multiplying network 203 at somefrequency f and some quality factor Q, then values for the lumpedelements 216 such as the inductances L and capacitances C or otherlumped elements are determined. The quality factor Q is definedconventionally as

Q=f/Δf.

Such values may be calculated from the known characteristic impedanceZ_(o) and the transmission line complex propagation constant γ of apredetermined portion of the hypothetical transmission line 101 (FIG. 3)of the hypothetical power multiplier 100 a. In this respect, thecharacteristic impedance Z_(o) and the transmission line complexpropagation constant γ may be calculated for a predefined unit length ofthe hypothetical transmission line 101 as follows:

Z=R _(T) +jωL _(T),

Y=G _(T) +jωC _(T),

Z _(o)=√{square root over (Z/Y)}=√{square root over ((R _(T) +jωL_(T))/G _(T) +jωC _(T)))}, and

γ=√{square root over (ZY)}=√{square root over ((R _(T) +jωL _(T))(G _(T)+jωC _(T)))}{square root over ((R _(T) +jωL _(T))(G _(T) +jωC _(T)))},

where Z is the series impedance per unit length of transmission line, Yis the shunt admittance per unit length of transmission line. In the lowloss case (i.e. R_(T)≈0 and G_(T)≈0), the characteristic impedancereduces to

Z _(o)√{square root over (L _(T) /C _(T))}.

In addition, the velocity of propagation may be calculated as

$v = {\frac{1}{\sqrt{L_{T}C_{T}}}.}$

In order to determine values for R_(T), L_(T), G_(T), and C_(T), for agiven section of transmission line 159, various references may beconsulted that provide such information such as, for example, Terman, F.E., Radio Engineering Handbook, McGraw-Hill, 1943, pp. 172-178, 191-215,or other references as can be appreciated.

Once the characteristic impedance Z_(o) for a predefined portion of thehypothetical transmission line 101 is known, then the complex electricallength θ of the predefined portion of the hypothetical transmission line101 is calculated as

θ=γl

where l is the physical length of the predefined portion of thehypothetical transmission line 101. Given the characteristic impedanceZ_(o), the transmission line complex propagation constant γ, and theelectrical length θ of the predefined portion of the hypotheticaltransmission line 101, the series impedances Z₁ and Z₂, and the shuntimpedance Z₃ of the T-network 183 (FIG. 6A) may be calculated asfollows:

Z ₁ =Z ₂ =Z _(o) tan h(θ/2), and

Z ₃ =Z _(o)/sin h(θ).

Alternatively, the shunt impedances Z_(A) and Z_(B), and the middleimpedance Z_(C) of the π-network 186 may be calculated as follows:

Z _(A) =Z _(B) =Z _(o) cot h(θ/2), and

Z _(C) =Z _(o) sin h(θ).

Once the series impedances Z₁ and Z₂, and the shunt impedance Z₃ of theT-network 183, or the shunt impedances Z_(A) and Z_(B), and the middleimpedance Z_(C) of the π-network 186 are known, then correspondingvalues for L and C may be determined. Assuming, for example, that onehas calculated the shunt impedances Z_(A) and Z_(B), and the middleimpedance Z_(C) of the π-network 186, then inductance L associated withthe middle impedance Z_(C) may be calculated therefrom where

Z _(C) =r+jωL.

Also, the capacitance C associated with the shunt impedances Z_(A) andZ_(B) may be calculated where

$Z_{A} = {Z_{B} = {\frac{1}{{j\omega}\; C}.}}$

It may be the case that L and C are too large to be practicallyrepresented in the form of a lumped element 216. If such is the case,then a reverse calculation or reverse mapping may be performed usingknown values for L and C to determine how much of the hypotheticaltransmission line 101 may be represented by a given T-network 183 orπ-network 186. In this respect, one may determine how many T-networks183 or π-networks 186 may necessarily be employed in a given powermultiplying network 203. In this respect, values may be chosen for L andC in view of the calculated values for L and C identified above.

Assuming that the series impedances Z₁ and Z₂, and the shunt impedanceZ₃ of the T-network 183 are calculated from predetermined values for Land C, then the characteristic impedance Z_(o) and the transmission linecomplex propagation constant γ may be calculated as follows:

${Z_{o} = \sqrt{Z_{1}( {Z_{1} + {2Z_{3}}} )}},{and}$$\gamma = {{{Arctanh}( \frac{\sqrt{Z_{1}( {Z_{1} + {2Z_{3}}} )}}{Z_{1} + Z_{3}} )}.}$

Alternatively, assuming that the shunt impedances Z_(A) and Z_(B), andthe middle impedance Z_(C) of the π-network 186 are calculated frompredetermined values for L and C, then the characteristic impedanceZ_(o) and the transmission line complex propagation constant γ may becalculated as follows:

${Z_{o} = {Z_{A}\sqrt{\frac{Z_{C}}{Z_{C} + {2Z_{A}}}}}},{and}$$\gamma = {{{Arctanh}( \frac{\sqrt{Z_{C}( {Z_{C} + {2Z_{A}}} )}}{Z_{A} + Z_{C}} )}.}$

Once the length l of the hypothetical transmission line 101 that isrepresented by a specified T-network 183 or π-network 186 is known, thenone can determine how many similar T-networks 183 or π-networks 186 areneeded to simulate the impedance of the entire hypothetical transmissionline 101. Thus, by performing the forward and reverse calculationsdescribed above, one can determine general values for the inductances Land capacitances C of the power multiplying network 203.

In addition, the power multiplying network 203 further comprises a phaseshifter 219. The phase shifter 219 comprises a circuit constructed fromlumped-elements that is combined in series with a portion of thedirectional coupler 209 to make up the inductance L of the specificsection within which the directional coupler 209 is located.

The power multiplying network 203 also includes a diverter 223 thatcouples the power multiplying network 203 to a load 226. The diverter223 is defined herein as an electrical element or circuit that may beemployed to divert or redirect all or a portion of a traveling wave fromthe power multiplying network 203 to the load 226. In this respect, thediverter 223 may comprise, for example, a switch, relay, solid stateswitch, plasma switch, or other device with like capability. Thediverter 223 may also be a circuit that presents an electric window thatis biased using a predefined control voltage or current to divert theenergy within a traveling wave to the load 226, depending upon the stateof the control voltage or current, etc.

During operation, the power source 213 is employed to launch an excitingtraveling wave in the launching network 206. The exciting traveling wavemay be, for example, a sinusoidal wave or other appropriate shape. Thedirectional coupler 209 couples at least a portion of the excitingtraveling wave from the launching network 206 into the power multiplyingnetwork 203, thereby resulting in a traveling wave that propagateswithin the power multiplying network 203. Given that the electricallength of the power multiplying network 203 is an integer multiple ofthe wavelength of the power source 213 and that the directional coupler209 is equal to ¼ of the wavelength of the power source 213, then thetraveling wave that propagates within the power multiplying network 203is continually reinforced by the portion of the exciting traveling wavethat is coupled into the power multiplying network 203. Also, thetraveling wave propagates in a single direction around the powermultiplying network 203. This results in power magnification M of thepower of the traveling wave by a predefined factor that may be manytimes greater than the power of the power source 213, depending upon thelosses and tolerances of the lumped-elements 216 and other factors.

Both the exciting traveling wave launched into the launching network 206and the traveling wave that propagates around the power multiplyingnetwork 203 may be AC power signals such as electrical power signalsgenerated at 50 Hertz, 60 Hertz, 400 Hertz, or any other power frequencyas can be found in the electrical generation systems in the UnitedStates and countries around the world. However, in any event, thefrequency of the exciting traveling wave, the traveling wave, and thepower source 213 may be any frequency possible, although they typicallycorrespond to frequencies with wavelengths for which the closed pathlength of the power multiplying network 203 is approximately 1/10 thewavelength or less of the traveling wave.

When the exciting traveling wave is applied to the launching network206, the power of the traveling wave continually increases with timeuntil it reaches a maximum power. The maximum power is reached when thelosses in the power multiplying network 203 plus the losses in thematching load R_(L) are equal to the power supplied by the power source213. When the maximum power is reached, the diverter 223 may be actuatedto direct the traveling wave from the power multiplying network 203 tothe electrical load 226. In a typical situation, it may take up toapproximately a dozen cycles to reach maximum power in the powermultiplying network 203, although it is possible that maximum power maybe reached in more or less cycles. Alternatively, the diverter 223 maybe actuated to direct the traveling wave from the power multiplyingnetwork 203 at any time deemed appropriate such as, for example, whenthe energy accumulated in the power multiplying network 203 reaches anypredefined threshold, etc.

The power multiplier 200 provides significant advantages in that itfacilitates real power multiplication at lower power frequencies such asthe operating frequencies of electrical power distribution systemsaround the world that operate, for example, at 50 Hertz, 60 Hertz, 400Hertz, or other low frequencies. The velocity inhibiting nature of thepower multiplying network 203 facilitates the creation of a powermultiplier 200 that can operate at such low power generation frequencieswith astonishing size reduction. That is to say, where prior theory mayhave taught that power multipliers operating at conventional powergeneration frequencies might have required a hypothetical waveguide thatextended for thousands of kilometers as discussed with reference to FIG.3, now the same can be created in a compact size that fits, for example,in a small room.

The velocity of propagation of the traveling wave through the powermultiplying network 203 relative to the velocity of a traveling wavethrough free space is described herein as the velocity factor. Thevelocity inhibiting nature of the power multiplying network 203 providesfor velocity factors that are on the order of 1/1,000,000, although evensmaller velocity factors may be achieved.

In addition, the power multiplier 200 may further include a number oflaunching networks 206, each launching network 206 being coupled to thepower multiplying network 203 by a directional coupler 209. Such aconfiguration would facilitate a corresponding increase in the rate atwhich the power of the traveling wave accumulates during operation ofthe power multiplier 200.

In an alternative embodiment, the traveling wave may be a solitary wavethat propagates around the power multiplying network 203. In order topropagate a solitary wave around the power multiplying network 203, thepower multiplying network 203 is constructed so as to include nonlinearelements such as, for example, diodes, transistors, or other activecomponents so as to be nonlinear and dispersive. Thus, nonlinearcomponents are defined herein as components that provide an outputhaving an amplitude that is not linearly proportional to the input ascan be appreciated by those with ordinary skill in the art. Byconstructing the power multiplying network 203 from a suitable networkof nonlinear elements and/or a combination of linear and nonlinearelements, a solitary wave may be propagated around the power multiplyingnetwork 203. In this respect, the power source 213 would be a pulsegenerator that generates and launches an exciting traveling wave intothe launching network 206. To achieve power multiplication, a solitaryexciting traveling wave would have to be spatially synchronized with thesolitary traveling wave. In addition, the launching network 206, thedirectional coupler 209, and the phase shifter 219 may be constructed toinclude elements that are nonlinear and dispersive in nature tofacilitate the propagation of solitary waves there through.

It should be appreciated that as the gain of the power multiplyingnetwork 203 increases, its quality factor Q rises and its bandwidth BWnarrows around the operating frequency. In one embodiment, this may be adesirable asset for a strictly monochromatic system. Should broaderbandwidths BW be desired, the electrical bandwidth BW of the powermultiplying network 203 may be tailored for the specific application.For example, low-loss power multiplying networks 203 with broader andcontrolled-shape passbands may be constructed following variouselectrical filter design. See for example, Matthaei, G. L., L. Young,and E. M. T. Jones, Microwave Filters, Impedance Matching Networks, andCoupling Structures, McGraw-Hill, 1964; and Fano, R. M., TheoreticalLimitations on Broadband Matching of Arbitrary Impedances, Journal ofthe Franklin Institute, Vol. 249, 1950, pp. 53-83 and 129-155.

In another embodiment, the power multiplier 200 as described above mayalso be constructed incorporating so called “Tracking-Filter” designtechniques such that the electrical passband of the power multiplier 200can be dynamic and automatically controlled to coherently trackfrequency and phase variations of the power source 213 while maintainingthe desired operational properties described above. In implementing apower multiplier 200 with a dynamic electrical passband, the frequencyof the power source 213 is monitored and compared with the resonantfrequency of the power multiplying network 203. An error signal may begenerated from such a comparison and employed in a feedback loop todynamically modify the ring component parameters such as thelumped-elements of the power multiplying network 203 to tune it to thespectral variations of the power source 213. In such case, thelumped-elements described above may be parametrically dynamic withvariable parameters as can be appreciated.

Referring next to FIG. 9, shown is a schematic that provides one exampleof the phase shifter 219 according to an aspect of the presentinvention. The phase shifter 219 comprises a T-network 183 a (FIG. 7A),although a π-network 186 a may be employed as well. In this respect, thephase shifter 219 includes series inductances L_(T) and a shuntcapacitance C_(T). In this respect, the phase shifter 219 is constructedfrom lumped-elements as part of the power multiplying network 203.

The series inductances L_(T) and the shunt capacitance C_(T) arespecified so as to result in a phase shift φ_(s). The series inductancesL_(T) and/or the shunt capacitance C_(T) (assuming that a T-network 183a is employed) may be variable so as to allow the phase shift φ_(s) tobe adjusted as necessary to compensate for any inaccuracies in the phaseshifts φ_(s) of each section and in the phase shift 8 of the directionalcoupler 209. This is done to ensure that the total phase shift presentedby the power multiplying network 203 is an integer multiple of 360degrees for the wavelength of the power source 213. The specificcalculations that are performed to determine the values of theinductances L_(T) and the shunt capacitance C_(T) will be discussed.

With reference to FIG. 10, shown is a schematic that illustrates anexample of the directional coupler 209 according to an aspect of thepresent invention. The directional coupler 209 comprises a number oflumped-elements. Such a directional coupler 209 ensures that thetraveling wave propagates in a single direction along the powermultiplying network 203 and to achieve the reinforcement of thetraveling wave with the portion of the exciting traveling wave thatpropagates through the launching network 206. The directional coupler209 includes the quarter wavelength delay circuits 233 and 236. Notethat the directional coupler 209 may also be depicted as including thematching load R_(L) (FIG. 8).

With the foregoing discussion of the power multiplying network 203, thedirectional coupler 209, and the phase shifter 219, the total phaseshift presented by the power multiplying network 203 may be determinedas follows:

φ_(PMW)=φ_(s)(N−1)+φ+θ,

where N is equal to the number of sections in the power multiplyingnetwork 203.

In addition, the diverter (FIG. 8) may be constructed in a mannersimilar to the directional coupler 209 in which the values of thecoupling capacitances are used to control the rate at which energyexists the power multiplying network 203.

With reference back to FIG. 8, once we have determined the values forthe inductances L and capacitances C per section of the power multiplier200 that comprises T-networks 183 (FIG. 6A) or π-Networks 186 (FIG. 6B),then actual power magnification that can be achieved by the resultingpower multiplier 200 given the values for the lumped-elements (i.e. theshunt capacitances C and the series inductances L) may be determined.Specifically, the lumped-elements are specified to achieve a predefinedphase shift per section at the predefined operating frequency.

The progression of calculations that is performed to determine thevalues for the lumped elements 216 such as the capacitances C andinductances L of the power multiplier 200 is now discussed. In thefollow calculations, the assumption is made that each section of thepower multiplying network 203 comprise π-networks 186 (FIG. 6B). Tobegin, the operating frequency f of the power multiplier 200 isspecified. Also, both the inductance L and capacitance C of each sectionof the power multiplying network 203 are specified based upon the valuesfor such elements identified above. In addition, a quality factor Q isspecified for the inductances L of each section of the power multiplyingnetwork 203. The frequency in terms of radians/sec is calculated as

ω=2πf radians/sec.

Also, the resistance in each of each inductance L is calculated as

$r = {\frac{\omega \; L}{Q}\mspace{14mu} {{Ohms}.}}$

Thereafter, the impedance Z_(C) is calculated as follows:

Z _(C) =r+iωL Ohms,

where “i” represents √{square root over (−1)} as is known by those withordinary skill in the art. Given the capacitances C specified above, theshunt impedances Z_(A) and Z_(B) are calculated as follows:

$Z_{A} = {Z_{B} = {\frac{1}{{\omega}\; C}\mspace{14mu} {{Ohms}.}}}$

Next, the characteristic impedance Z₀ is calculated as follows:

$Z_{0} = {Z_{A}\sqrt{\frac{Z_{C}}{( {Z_{C} + {2Z_{A}}} )}}\mspace{14mu} {{Ohms}.}}$

The characteristic impedance is defined as the ratio of the forward wavevoltage over the forward wave current. In this respect, a physicalmeasurement of the characteristic impedance of each section may be takenand compared with the calculated characteristic impedance Z₀ to verifythe accuracy thereof.

In addition, the propagation constant γ per section is calculated asfollows:

$\gamma = {{{atanh}\lbrack \frac{\sqrt{Z_{C}( {Z_{C} + {2Z_{A}}} )}}{( {Z_{A} + Z_{C}} )} \rbrack}.}$

The Attenuation Constant α per section and the Phase Constant β persection are defined as

α_(section) =Re(γ)Nepers/section, and

β_(section) =Im(γ)radians/section.

The phase shift per section may then be calculated as

φ=(57.296 Deg/Rad)β_(section)Degrees.

The velocity of the traveling wave in sections per second propagatingalong the power multiplying network 203 is calculated as

$v = {\frac{\omega}{\beta_{section}}\mspace{14mu} {section}\text{/}{{second}.}}$

Next, the electrical circumference C_(Λ) of the power multiplyingnetwork 203 is specified in terms of wavelengths at the operatingfrequency in degrees as

C _(Deg) =C _(Λ)(360 Degrees/wavelength)Degrees.

Next, the number of sections N (either T-networks or π-networks) iscalculated as

$N = {\frac{C_{Deg}}{\varphi}.}$

Once the number of sections N is known, then the loss resistance R_(C)around the closed path of the power multiplying network 203 may becalculated as

R _(C) =Nr Ohms.

where r is as defined above. The field propagation decay A for a singletraversal of the power multiplying network 203 may be calculated as

A=e ^(−α) _(section) ^(N).

The attenuation A_(dB) around the power multiplying network 203 iscalculated as

A _(dB)=−20 log(A).

The pulse duration T of a peripheral disturbance is calculated as

$\tau = {\frac{N}{v}\mspace{14mu} {{seconds}.}}$

The power magnification M of the power multiplier 200 at optimumcoupling is calculated as

$M = {\frac{1}{( {1 - A^{2}} )}.}$

The power magnification M_(dB) expressed in decibels is calculated as

M _(dB)=10 log(M).

The optimum coupling C_(opt) is calculated as

C _(Opt)=1−A ².

The optimum coupling C_(opt) is calculated in decibels (dB) as

C _(optdB)=10 log(C _(opt))dB.

In addition, a useful reference that may be consulted to determine thevarious elements of the directional coupler 209 and the phase shifter219 is Matthaei, G. L., L. Young, and E. M. T. Jones, Microwave Filters,Impedance Matching Networks, and Coupling Structures, McGraw-Hill, 1964,(see Chapter 14). While specific circuit designs may be discussed hereinthat may be employed as the directional coupler 209 and the phaseshifter 219, it is understood that other circuit designs and circuitstructures may be employed as well, such alternative designs fallingwithin the scope of the present invention.

Power Multiplication and Parametric Excitation

With reference to FIGS. 11 and 12, shown are drawings of powermultipliers 400 a and 400 b that employ parametric excitation accordingto various embodiments of the present invention. The power multipliers400 a and 400 b each include a respective power multiplying network 403a and 403 b and a launching network 406. Each of the power multiplyingnetworks 403 a/403 b comprises a ring as mentioned above. The launchingnetwork 406 also includes a directional coupler 409 that couples thelaunching network 406 to the respective power multiplying networks 403a/403 b. A power source 413 is coupled to the launching network 406.Also, the launching network 406 is terminated in a matching load R_(L).

In addition, the each of the power multiplying networks 403 a/403 bfurther comprises a phase shifter 419. The phase shifter 419 comprises acircuit constructed from lumped-elements that is combined in series witha portion of the directional coupler 409 to make up the inductanceL(t)/L of the specific section within which the directional coupler 409is located.

Each of the power multiplying networks 403 a/403 b also includes adiverter 423 that couples the respective power multiplying network 403a/403 b to a load 426. The diverter 423 is defined herein as anelectrical element or circuit that may be employed to divert or redirectall or a portion of a traveling wave from one of the power multiplyingnetworks 403 a/403 b to the load 426. In this respect, the diverter 423may comprise, for example, a switch, relay, solid state switch, plasmaswitch, or other device with like capability. The diverter 423 may alsobe a circuit that presents an electric window that is biased using apredefined control voltage or current to divert the energy within atraveling wave to the load 426, depending upon the state of the controlvoltage or current, etc.

In the embodiment shown in FIGS. 11 and 12, each of the powermultiplying networks 403 a/403 b is constructed from reactances 416 andparametric reactances 418 that, according to one embodiment, compriselumped-elements. As shown in FIG. 11, the reactances 416 comprisecapacitances C (FIG. 14) and inductances L (FIG. 15) and the parametricreactances 418 comprise parametric inductances L(t) (FIG. 14) andparametric capacitances C(t). Alternatively, the parametric reactances418 in a single power multiplier may include both parametric inductancesL(t) and parametric capacitances C(t).

In one embodiment, the parametric reactances 418 such as the parametricinductances L(t) or the parametric capacitances C(t) may comprise linearor non-linear reactances. Thus, the parametric inductances L(t) maycomprise linear or non-linear inductances and the parametriccapacitances C(t) may comprise linear or non-linear capacitances. Asdescribed herein, a linear circuit component is one in which theimpedance of the circuit component is not a function of the magnitude ofa voltage signal in the circuit component.

Examples of linear reactances may comprise, for example, conventionalair-core coils of wire, capacitors, and similar lossless passive circuitelements composed of media with linear permeability (μ) and lineardielectric permittivity (∈). Sections of transmission lines (andwaveguides) constructed of linear media may also supply examples oflinear reactances as viewed at their inputs.

Examples of non-linear reactances may comprise reactive elements with anonlinear impedance relationship between voltage and current. Suchreactive elements may include saturable reactors, varactor (varicap)diodes with voltage variable junction capacitance, and elementsconstructed of nonlinear permittivity [∈=∈(V), where permittivity is afunction of voltage] or nonlinear permeability [μ=μ(I), wherepermeability is a function of current], as is the case in transmissionline elements with magnetically biased ferrites, and even plasma media.

Time-varying reactive elements also occur in engineering practice.Common examples of time-varying reactances are inductors and capacitorswhose permittivity and permeability functions are pumped in time by acontrol voltage or current. Similarly, distributed time-varyingimpedances have their constitutive parameters pumped by a controlsignal, which may be electrical, electromagnetic, optical, thermal,mechanical, acoustical, etc.

The power multipliers 400 a and 400 b are operated in order to multiplypower in much the same way as the power multiplier 200 (FIG. 8) with theexception that the parametric reactances 418 are varied in time as willbe described below. As a result of the variation of the parametricreactances 418, a negative resistance is introduced into the powermultiplying networks 403 a/403 b that effectively electrically negatesthe physical resistance inherent in the components of the powermultiplying networks 403 a/403 b. Consequently, the power multipliers400 a/400 b can accumulate a drastically greater amount of power in atraveling wave within the power multiplying networks 403 a/403 b. In thecase that the physical resistance of a respective power multiplyingnetwork 403 a/403 b is almost completely negated, then the powermultiplying network 403 a/403 b may actually approach superconductivity.

Recall as described above that in a power multiplier 200 (FIG. 8) thatdoes not employ parametric excitation (does not make use of parametricreactances), power will continue to build up in the power multiplyingnetwork 203 (FIG. 8) (or the ring) during operation until the losses dueto the inherent resistance of the power multiplying network 203 plus thelosses in the matching load R_(L) that terminates the launching network406 is equal to the power generated by the power source 413. Given theuse of parametric reactances 418 in the power multiplying networks 403a/403 b, the physical resistance of the power multiplying networks 403a/403 b are negated by the negative resistance introduced due to theparametric excitation of the parametric reactances.

Thus, if the physical resistance was reduced due to the negativeresistance injected in the power multiplying networks 403 a/403 b, thepower of the traveling wave in the power multiplying networks 403 a/403b will continue to build up until the losses due to the reduced physicalresistance and due to the matching load equal the power generated by thepower source 413. As the negative resistance injected into a powermultiplying network 403 a/403 b approaches the total physical resistanceof a respective one of the power multiplying networks 403 a/403 b, thenthe power multiplying networks 403 a/403 b approach superconductivity.However, it may be the case that there are limits to how closely thenegative resistance can approach the actual physical resistance of therespective power multiplying network 403 a/403 b, where the actualamount of negative resistance depends upon the magnitude and phase ofthe variation in the parametric reactances that make up part of thepower multiplying network 403 a/403 b. Thus, the actual amount ofnegative resistance generated is application specific.

According to the various embodiments, one or more parametric reactances418 in the power multiplying networks 403 a/403 b are varied in time ata frequency that is in a predefined relationship relative to theoperating frequency of the power source 413. That is to say, thefrequency of at which the parametric reactances 418 are varied in timeis in a predefined relationship relative to the frequency of a travelingwave in the ring of the power multiplier 400 a/400 b.

To explain further, if a signal at frequency f_(s) is injected into alinear, tuned circuit such as an LC circuit, which has a reactiveelement changing at frequency f_(p) (called the “pump” frequency), a newmixer frequency (called the idler frequency, f_(i)) will appear in thecircuit. The relation between these three frequencies is as follows:

f _(i) =mf _(p) ±nf _(s).

If the reactance element is varied at a frequency f_(p) (the pumpfrequency) in a 2:1 ratio to the resonant frequency (which is also tunedto the signal frequency f_(s)) of the circuit, then the difference oridler frequency f_(i) will be the same as the signal frequency f_(s).

If the operating point of the reactance element is varied at onefrequency, and an oscillator signal is coupled into the circuit atanother frequency, under certain conditions between the two frequencies,the circuit impedance, instead of being a pure imaginary, will becomecomplex. The imaginary component could correspond to an inductivereactance, but a negative real part can arise, effectively correspondingto a negative resistance injected into the circuit. The amount ofnegative resistance injected into the circuit is controlled by therelative magnitude and phase at which the reactance element is varied.

According to various embodiments, the negative resistance arises whenthe pump frequency f_(p) is equal to twice the signal frequency f_(s) ofthe circuit. In this “degenerate” mode of operation, where both m andn=1 and f_(p)=2×f_(s), the idler frequency f_(i) is equal to the signalfrequency f_(s). Therefore, when the pump frequency f_(p) is equal totwice the signal frequency (2×f_(s)), a negative resistance is injectedinto the circuit.

According to the various embodiments of the present invention, which, bydesign, possess small dissipation, the parametric reactances 418 arevaried at a frequency that is twice the frequency of the power source413 (and the generated traveling wave flowing through the ring) of thepower multiplier 400 a/400 b. When this condition is met, the parametricreactance effectively negates at least a portion of the physicalresistance inherent in the power multiplying networks 403 a/403 b. Thedegree to which the physical resistance inherent in the powermultiplying networks 403 a/403 b is negated depends upon the magnitudeof the negative resistance generated by the operation of the parametricreactances. The magnitude of the negative resistance depends upon themagnitude and phase of the variation of the parametric reactances 418and is design specific as described above.

While, in one embodiment, a ratio of 2:1 is specified between thefrequency of variation of the parametric reactances 418 and thefrequency of the power source 413 (or traveling wave in the ring) inorder to generate the negative resistance in the power multiplyingnetworks 403 a/403 b, it is understood that other frequencyrelationships between the frequency of variation of the parametricreactances 418 and the frequency of the power source 413 may bespecified in order to generate the negative resistance in the powermultiplying networks 403 a/403 b. For example, to aid in determiningsuch other frequency relationships between the frequency of variation ofthe parametric reactances 418 and the frequency of the power source 413,see the topic of Mathieu's Equation in Cunningham, W. J., Introductionto Nonlinear Analysis, McGraw-Hill, 1958, pp. 259-280) on circuitdissipation.

As Cunningham demonstrates, growing oscillations (negative resistance)and regions of instability may occur in second order systems for certainother noninteger (i.e. not commensurable) frequency ratios, which, inthe presence of damping, also depend upon system dissipation as well asthe amplitude and phase of the parameter variation. These are commonlymanifested as the region of instability “tongues” in the solutions ofMathieu's equation. Also, see Kharkevich, A. A., Nonlinear andParametric Phenomena in Radio Engineering, John F. Ryder Publisher,Inc., 1962, pp. 166-166; Mandelshtam, L. I. and Papaleksi, N. D., “Onthe Parametric Excitation of Electric Oscillations,” Soviet Journal ofTechnical Physics, Vol. 4, No. 1, 1934, pp. 5-29. See FIG. 1, p. 9.)Such other frequencies may also be employed, although efficiency maysuffer at such other ratios.

The parametric reactances 418 may be implemented using any one of anumber approaches. In particular, the parametric reactances 418 may becreated electrically, mechanically, thermally, or via some otherapproach. Specific examples include the creation of a parametricinductance L(t) using a magnetic amplifier that involves varying thepermeability μ of a core about which an inductor is wound. As thepermeability μ is altered, so is the inductance of the winding. Also, inanother example, the parametric capacitance C(t) may be created using adielectric amplifier in which the permittivity £ of a dielectricassociated with the parametric capacitance C(t) is varied over time. Asthe permittivity ∈ is varied, the resulting capacitance of theparametric capacitance C(t) is varied. Also, the parametric reactancemay be created mechanically using wheels and the like. For a moredetailed discussion as to the various devices that may be used orapproaches taken to implement the parametric reactances 418, referenceis made to the following papers:

-   Manley, J. M. and Peterson, E., “Negative Resistance Effects in    Saturable Reactor Circuits,” AIEE Transactions, Vol. 65, December    1946, pp. 870-88;-   Mandelstam, L., Papalexi, N., Andronov, A., Chaikin, S., and Witt,    A., “Report on Recent Research on Nonlinear Oscillations,” Technical    Physics of the USSR, Leningrad, Volume 2, Number 2-3, pp 81-134,    1935;-   Mumford, W. W., “Some Notes on the History of Parametric    Transducers,” Proceedings of the IRE, Vol. 48, May 1960, pp.    848-853; and-   Raskin, J-P., Brown, A. R., Khuri-Yakub, B. T., Rebeiz, G. M., “A    Novel Parametric-Effect MEMS Amplifier,” Journal of    Microelectromechanical Systems, Vol. 9, December 2000, pp. 528-537.    Each of these references is incorporated herein by reference in    their entirety.

According to one embodiment, the power multipliers 400 a and 400 b areeach described as including a power multiplying network 403 a/403 bcomprising a multiply-connected, velocity inhibiting circuit constructedfrom a number of lumped-elements.

In addition, the term “lumped” is defined herein as effectivelyconcentrated at a single location. Thus, the terms “lumped-elements”refer to discrete, two-terminal, concentrated electrical elements suchas capacitance, inductances, resistance, and/or conductance. Thus, thelumped-elements as described herein may comprise discrete inductors,capacitors, or resistors. In addition, as contemplated herein,lumped-elements may also comprise diodes, transistors, and othersemi-conductors that may be described, for example, as nonlinearresistors or conductors that have resistance or conductance that iscontrolled by the polarity of applied voltages or currents, etc. Inaddition, lumped-elements may also comprise inherent capacitances,inductances, resistances, or conductances of various electricalstructures such as helices, parallel plates, or other structure as willbe discussed. Similar to the power multiplying network 203, thedirectional coupler 209 is also constructed using lumped-elements.

Lumped elements are circuit components for which the parametersinductance, capacitance, resistance, and conductance have the samemeaning as in the static situation. That is to say, the first orderterms in the electromagnetic power series solution of Maxwell'sequations are adequate to describe “lumped element” networks. Thisusually occurs when the dimensions of the components, includinginterconnections, is very small compared to the wavelength of operation.As the ratio of network dimension to wavelength is decreased, Maxwell'srigorous field equations transition to the distributed-elementtransmission line equations of Heaviside, and, in the limit, these passto the lumped element circuit equations of Kirchoff. Only the latter arenecessary to describe electrical circuits operating in the lumpedelement regiem of conventional electronic circuit theory. No spatialintegrals or spatial derivatives are necessary to analyze discretecomponent networks as would be required for distributed-elements andradiating circuits.

According to various embodiments, the parametric excitation as describedabove applies to power multipliers that comprise distributed elementcircuits as well. A distributed element circuit is a circuit whosereactance (inductance and/or capacitance) are distributed over aphysical distance that is comparable to an operating wavelength of thecircuit. Power multipliers that include a ring comprising a distributedelement circuit include those that substantially are not velocityinhibiting circuits. For distributed element power multipliers,Maxwell's rigorous field equations transition to the distributed-elementtransmission line equations of Heaviside. A distributed element powermultiplier are analyzed using spatial integrals or spatial derivatives.A distributed element power multiplier is one with a waveguidecomprising a ring with a physical circumference that is an integermultiple of the wavelength λ_(w) of an exciting traveling wave thataccumulates in the ring. It follows that for a distributed element powermultiplier to be practical, the frequency of operation is relativelyhigh, resulting a relatively small structure that can be practicallyconstructed. In this respect, the power multiplier 100 (FIG. 1)comprises a distributed element power multiplier.

In another embodiment, the principles of parametric excitation of apower multiplier described above with reference to the lumped-elementpower multipliers 400 a/400 b apply equally to distributed element powermultipliers. The reactance parameters of distributed ring powermultipliers may be varied in space and time (“pumped”) appropriately byany one of a variety of means. For example, the transmission lineelectrical parameters may be pumped by employing a magnetically biasedplasma. They may be pumped by using a magnetically biased permeabilityvarying in space and time, μ(x,t). Also, they may be pumpedappropriately by using a voltage-controlled electrical permittivity,∈(x,t), in the transmission line. Such techniques, obviously, have theeffect of pumping the inductance per unit length L(x,t) and thecapacitance per unit length C(x,t), respectively, at the desired pumpfrequency and phase. For one discussion of techniques for pumpingreactance variations in the context of conventional distributed elementparametric amplifiers, reference is made to Louisell, W. H., CoupledMode and Parametric Electronics, John Wiley and Sons, Inc., 1960, pp.131-147, and the references cited and discussed therein, which isincorporated herein by reference.

The power multipliers described (i.e. the power multipliers 400 a/400 b,and other embodiments such as distributed element power multipliers)above that employ the principles of parametric excitation may beemployed in the same manner as the power multipliers 200 and 250described above. For example, the power multipliers 400 a/400 b may beemployed for power smoothing with respect to power distribution networks300 as described above. Also, the power multipliers 400 a/400 b (ordistributed element power multipliers employing parametric excitation)may be employed in any other situation where the multiplication orstorage of power is desired, where the essential difference between thepower multipliers that employ parametric excitation and the powermultipliers that do not is that the power multipliers that employparametric excitation can buildup or store a vastly greater amount ofpower due to the fact that the physical resistance in the ringsassociated with such power multipliers is negated as described above.

Power Processing

With reference to FIG. 13, shown is a multiply-connected power processor500 according to various embodiments. The multiply-connected powerprocessor 500 includes a power processing network 503. According to oneembodiment, the power processing network 503 is a multiply-connectedvelocity inhibiting circuit. The power processing network 503 includesparallel capacitors C and active circuits 506 that are coupled in seriesaround the power processing network 503 as shown. The active circuits506 synthesize a passive lumped element in the power processing network503. In particular, the active circuits 506 are configured to synthesizean inductance such as, for example, the inductances L as describedabove, for example, with reference to FIG. 8.

An alternating current (AC) power source 509 is coupled to the powerprocessing network 503 by a source coupling 513. The AC power source 509may be, for example, one or more generators on a power grid or othergenerator. Such AC power sources 509 typically operate at powerfrequencies such as 50 Hz., 60 Hz., 400 Hz., or other power frequency.Alternatively, the AC power source 509 may be some other type of sourcethat can operate at virtually any frequency for which the powerprocessing network 503 is designed.

The source coupling 513 comprises, for example, a directional coupler aswill be described below. A phase shifter 516 is included in the powerprocessing network 503 in series with the source coupling 513. The phaseshifter 516 provides for adjustment of the total impedance presented bythe source coupling 513 relative to the rest of the power processingnetwork 503 to ensure that the electrical length of the power processingnetwork 503 is a multiple of a wavelength of a resonant frequency of thepower processing network 503.

Coupled to each of the active circuits 506 is a power source V_(S) thatprovides power to the active circuit 506 so that it may properlyfunction. The power source V_(S) may be, for example, a direct current(DC) power source or an alternating current (AC) power source. In oneembodiment, the active circuit 506 may include rectification circuitryin order to convert an AC power source to DC power in order to provideDC power to various components included therein. The active circuit 506may employ various types of circuit components such as, for example,operational amplifiers or other active circuitry as will be described.

An electrical load 523 is coupled to the power processing network 503 byway of a load coupling 526. The load coupling 526 may comprise, forexample, a reverse directional coupler as will be described below. Inaddition, a phase shifter 527 is included in the power processingnetwork 503 in series with the load coupling 526 to adjust the totalimpedance presented by the load coupling 526 and the phase shifter 527in conjunction with the design of the power processing network 503. Thesource coupling 513 and the load coupling 526 each may be controlled soas to vary the amount of power that they couple into and out of thepower processing network 503 at any given time as will be described.

Where the power sources V_(S) are direct current power sources, they maycomprise any one of a number of different types of direct current powersources such as, for example, batteries, solar cells, rectified ACsources, or other DC sources as can be appreciated. Also, where thepower sources V_(S) comprise AC power sources, such may compriseconventional utility power sources, AC power generated by wind farms,water current generation systems, or other generators as can beappreciated. Such generators may be may be nuclear reaction based,solar, chemical and electrochemical, thermal (including geothermal,plasma, as well as combustion), and mechanical energy conversiondevices.

The multiply-connected power processor 500 further comprises a controlsystem 529 that controls the operation of the source coupling 513 andthe load coupling 526 based upon an input from the electrical load 523as will be described. The control system 529 may comprise a digital oranalog control system as can be appreciated. In one embodiment, thecontrol system 529 employs a processor circuit or other type of digitalcircuitry.

In addition, a backup power source 531 may be coupled to the powerprocessing network 503 using a source coupling 513 with a correspondingphase shifter 516. The backup power source 531 may comprise, forexample, a backup generator or other AC power source. Further, backupbatteries may be employed where an AC power signal is generatedtherefrom using a DC to AC converter as can be appreciated.

Next, a general discussion is provided as to the operation of themultiply-connected power processor 500 according to the variousembodiments. To begin, the AC power generation source 509 generates ACpower that is applied to the power processing network 503 through thesource coupling 513. The electrical length of the power processingnetwork 503 is designed to be a multiple of the wavelength of the signalgenerated by the AC generation source 509. As a consequence, the powerprocessing network 503 resonates at the frequency of the AC generationsource 509. The active circuits 506 effectively synthesize or emulatethe passive inductances as described above. Where the multiply-connectedpower processor 500 is designed to resonate at low frequencies such aspower frequencies (i.e. 50 Hz, 60 Hz, or 400 Hz), the active circuits506 may be much smaller in size than the physical passive inductancesthey emulate.

The multiply-connected power processor 500 processes the power generatedby the AC generation source 509 that enters the power processing network503. In this respect, the multiply-connected power processor 500 caninput unprocessed power through the source coupling 513 and outputs“clean” AC power to the load 523 through the load coupling 526.Unprocessed power is power that is subject to various anomalies such astransients, harmonics, voltage surges, voltage sags, brownouts, or otherproblems. Once the power is coupled into the multiply-connected powerprocessor 500, the power is processed by virtue of the characteristicsand operation of the multiply-connected power processor 500 to minimizeor eliminate any of the above anomalies so as to output clean power tothe load 523.

To explain further, the power generated by the AC power generationsource 509 is coupled into the power processing network 503 and travelsaround the power processing network 503 as a traveling wave at theresonant frequency of the power processing network 503 as describedabove. The active circuits 506 emulate or synthesize inductances in thepower processing network 503 such that the power processing network 503has an electrical length that is a multiple of the wavelength of thesignal generated by the AC power generation source 509. The controlsystem 529 controls the load coupling 526 to divert power out of thepower processing network 503 to the electrical load 523 as dictated bythe power demand of the electrical load 523 at any given instant.

The control system 529 also controls the source coupling 513 to controlthe amount of power that enters the power processing network 503 at anygiven instant. The source coupling 513 can be manipulated by the controlsystem 529 to input power at a rate that is equal to the rate at whichpower is diverted out of the ring by the load coupling 526. Also, thecontrol system 529 may manipulate the source coupling 513 to input powerinto the power processing network 503 to maintain an elastic reservoirof power at any given instant. Specifically, the control system 529 maybe configured to maintain or store a predefined amount of power in thepower processing network 503. The actual power storage capacity of thepower processing network 503 depends upon the power rating or capacityof the active circuits 506. The energy stored in the power processingnetwork 503 may be input into the power processing network 503, forexample, at initial startup when first applying power to the electricalload 523 or at any other time. To build up excess energy in the powerprocessing network 503, the control system 529 controls the sourcecoupling 513 to input more power into the power processing network 503than is routed to the electrical load 523 through the load coupling 526.To this end, a gain is associated with the operation of the powerprocessing network 503 to the extent that power can build up in the formof the traveling wave.

Once a desired amount of power is stored in the power processing network503, when greater amounts of power are demanded by the electrical load523, the control system 529 may be configured to respond by manipulatingthe load coupling 526 to divert a greater amount of power from the powerprocessing network 503 to the electrical load 523. At the same time, thesource coupling 513 may be manipulated to increase the power input intothe power processing network 503 by the same amount so that the amountof power stored in the power processing network 503 remainssubstantially constant.

The power from the AC power source 509 may experience various anomaliesfrom time to time. For example, it may be the case that voltage sags orbrownouts occur due to various activity on the power grid to which theAC power source 509 is coupled. A voltage sag is a drop in the magnitudeof the power signal below nominal values usually for a short period oftime measured in several milliseconds, although a voltage sag may haveany time duration. A brownout is a short term loss of power usuallymeasured in terms of several cycles of the power signal. Themultiply-connected power processor 500 protects the electrical load 523from voltage sags and brownouts by storing excess power in the powerprocessing network 503 that may be diverted from the power processingnetwork 503 to the load 523 to compensate for the loss of powerrepresented by a voltage sag or brownout. Given that the power capacityof the active circuits 506 may limit the amount of power that can bestored in the power processing network 503, the capacity of the powerprocessing network 503 to compensate for a voltage sag or brownout islimited by the amount of power that can be stored in the powerprocessing network 503.

Also, the power processing network 503 may be employed to absorb voltagesurges or overvoltages, thereby preventing such voltage surges orovervoltages from reaching the electrical load 523 and potentiallycausing physical damage. Typically, voltage surges or overvoltages maylast several cycles of the power signal, although it is possible theymay last for much longer. To this end, the control system 529 may beconfigured to maintain the power in the power processing network 503below full capacity in order to be able to absorb voltage surges orovervoltages by storing the excess energy in the power processingnetwork 503.

In addition, the control system 529 receives an input indicating themagnitude of the power voltage from the AC power source 509. Based uponthis input, the control system 529 may manipulate the source coupling513 to input a lesser amount of power into the power processing networkduring an occurrence of a voltage surge or overvoltage. Thus, the powerprocessing network 503 may be configured to absorb an initial amount ofexcess energy from a voltage surge until the control system 529 canreact to reduce the rate at which power is input into the powerprocessing network 503. When the voltage surge or overvoltage is overand the power voltage from the AC power source 509 has returned tonominal, the control system 529 can adjust the source coupling 513 toreestablish the rate of power input that existed before the voltagesurge or overvoltage. The ability of the power processing network 503 tostore energy in the form of the traveling wave provides a degree offlexibility in responding to anomalies such as voltage surges orovervoltages.

Of course, while adjustments are made to the source coupling 513resulting in a change in the amount of power stored in the powerprocessing network 503, the load coupling 526 is adjusted to ensure thatthe amount of power diverted to the electrical load 523 matches thepower demand of the electrical load 523 at any given instant. Thus, thecontrol system 529 controls the source coupling 513 based upon detectionof anomalies such as voltage surges or overvoltages, and based upon thepower demand of the electrical load 523 at any given instant.Ultimately, the reservoir of power stored in the power processingnetwork 503 provides time to allow the control system 529 to react topower input problems while maintaining a clean, robust power output tothe electrical load 526.

In addition, by virtue of the resonant nature of the power processingnetwork 503, the power processing network 503 acts as a narrow band passfilter that will filter out all voltage transients or other anomalies inthe signal generated by the AC power generation source 509. Voltagetransients are aberrations such as high frequency waveforms imposed ontop of a power voltage. Voltage transients may be created by variousoperational activity on an electrical grid or by natural phenomena likelightning. Typical transients have a frequency that differs from thenominal frequency of the power voltage. For example, a voltage transientmay comprise a ringwave with oscillating frequencies as high as 100 KHzor more.

Due to the fact that the power processing network 503 acts as anarrowband pass filter at the nominal frequency of the power voltage,such voltage transients or other anomalies are either dissipated in theform of heat in the components of the multiply-connected power processor500 or are converted into the power signal itself. Ultimately, suchunwanted aberrations in the power voltage are prevented from reachingthe load 523. In one embodiment, the active circuits 506 employ thepower source V_(S) to overcome any inherent resistance in such activecircuits 506 such that they can emulate an ideal inductance without anyresistive loss. Also, voltage clamping technologies or othertechnologies may be employed at the input of the source coupling 513.Such technologies may comprise, for example, metal oxide varistors,zener diodes, gas tubes, or other components that may be employed toclamp the high voltages of transients to acceptable levels before theyenter the power processing network 503 to reduce the possibility ofdamage to the components of the power processing network 503 itself.

In addition, the multiply-connected power processor 500 may be employedto ensure an uninterruptible supply of power to the electrical load 523due to the backup power source 531 coupled to the power processingnetwork 503 through the additional source coupling 513. Upon a loss ofthe AC power signal from the AC power source 509, the control system 529may react by manipulating the additional source coupling 513 to inputpower into the power processing network 503 from the backup power source531. Also, the control system 529 may be configured to start up thebackup power source 531 when necessary such as might be the case with abackup generator.

In cases where the backup power source 531 needs time to be broughtonline such as might be the case with a backup generator, the electricalload 523 may be supplied with power from the excess power stored in thepower processing network 503. Thus, the AC power storage characteristicsof the power processing network 503 can be employed to provide a smoothtransition between alternative power sources coupled to the powerprocessing network 503, thereby ensuring that the electrical load 523never sees an interruption in the power supplied thereto. This is usefulfor critical loads as can be appreciated.

When the backup power source 531 is first brought up, it is desirablethat the phase and frequency of the power signal generated thereby matchthe phase and frequency of the traveling wave that travels around thepower processing network 503. To this end, the phase of the power signalmay be dynamically shifted as needed so as to match the phase of thetraveling wave in the power processing network 503.

The active circuits 506 may comprise, for example a gyrator circuit orother circuit that facilitates the emulation of the inductances L asdescribed above. In one embodiment, the gyrator circuits employed as theactive circuit 506 may synthesize a floating inductance L as will bedescribed. The active circuits 506 may also synthesize a non-floatinginductance. Also, the active circuits 506 may comprise a variableamplifier gain that is controlled by the control system 529 in a manneras will be described below. If the active circuits 506 include variableamplifier gain, then they may be manipulated to add power to the powermultiplying circuit from the power source V_(S) in a manner as will bedescribed below.

Referring next to FIG. 14A, shown is a schematic of a gyrator circuit506 a that emulates a floating inductance as can be appreciated. Anideal gyrator is a lossless, passive two-port component with a realconstant k called the gyration ratio. A gyrator circuit may beconfigured to invert an impedance such that it can make a capacitivecircuit such as a capacitor behave inductively. Also, a gyrator maycause a band pass filter to behave much like a band stop filter and soon. As shown in FIG. 14A, two gyrators 533 are coupled to a commoncapacitor C, where two of the leads of the respective gyrators 533 arecoupled together internally, thereby forming the gyrator circuit 506 a.

As seen with respect to FIG. 14B, the ultimate result is a circuit thatemulates a floating inductance where the two bottom leads are coupledtogether internally and are not coupled to external devices. Theultimate inductance L of this circuit is calculated as L=Ck². Thus, thefloating inductance may be designed by properly specifying the gyrators533 and the capacitance C to provide for a desired inductance L to beused in the power processing network 503.

Thus, the gyrator circuit 506 a of FIG. 14A is one example of an activecircuit 506 that may be employed in the power processing network 503according to the various embodiments. While the one example is describedwith respect to FIGS. 14A and 14B, it is further understood that thereare many other different types of active circuits that may be employedto emulate an inductance as described with respect to the powerprocessing network 503 of FIG. 13.

It should be noted that the floating inductance presented by the gyratorcircuit 506 a may be varied dynamically by varying the capacitance C. Assuch, the gyrator circuit 506 a may be employed as a parametricreactance as described above.

Referring back to FIG. 13, in one embodiment, each of the activecircuits 506 comprises a gyrator circuit 506 a (FIG. 14A) thatsynthesizes a floating inductance as described above. In addition, theinductances synthesized by the active circuits 506 may be variedparametrically in order to generate a negative resistance in the powerprocessing network 503 as was described above with reference to FIGS. 11and 12 above. For example, the inductance synthesized by the gyratorcircuit 506 a (FIG. 14A) may be varied by varying the capacitance C(FIG. 14A).

Also, it should be noted that other circuits beyond the gyrator circuit506 a may be employed as the active circuits 506 to synthesize theinductances of the power processing network 503. In particular, suchcircuits may comprise, for example, circuits that synthesizenon-floating inductances, etc.

Referring next to FIG. 15, shown is one example of the source coupling513 according to various embodiments. The source coupling 513 includescoupling capacitors C_(P1) and C_(P2). Also, the source coupling 513includes quarter wavelength delay circuits 536 and 539. The delaycircuits 536 and 539 provide for a quarter wavelength delay in the powersignal received from the AC power source 509 as well as the power signalon the power processing network 503. By virtue of the quarter wavelengthdelay circuits 536 and 539, and the coupling capacitors C_(P1) andC_(P2), at least a portion of the power signal generated by the AC powersource 509 is coupled into the power processing network 503 thatpropagates around the power processing network 503 as described above.The coupling capacitors C_(P1) and C_(P2) are variable capacitors inorder to control the amount of power from the AC power source 509 thatis actually coupled into the power processing network 503. The activecircuits 506 synthesize inductances in each of the quarter wavelengthdelay circuits 536 and 539 as shown. The control system 529 controls therate at which power is coupled into the power processing network 503 bycontrolling the values of the coupling capacitors C_(P1) and C_(P2).

With reference then to FIG. 16, shown is one example of a load coupling526 according to various embodiments. The load coupling 526 includescoupling capacitors C_(L1) and C_(L2). Also, the load coupling 526includes quarter wavelength delay circuits 543 and 546. The delaycircuits 543 and 546 provide for a quarter wavelength delay in the powersignal traveling in the power processing network 503 and in a divertedpower signal applied to the electrical load 523. The coupling capacitorsC_(L1) and C_(L2) may be variable so as to control the amount of powerapplied to the electrical load 523 at any given instant. The controlsystem 529 is configured to dynamically control the capacitance of thecoupling capacitors C_(L1) and C_(L2) so as to control the amount ofpower directed to the electrical load 523 based on feedback from theelectrical load 523. By virtue of the quarter wavelength delay circuits543 and 546, and the coupling capacitors C_(L1) and C_(L2), at least aportion of the power signal traveling in the power processing network503 is coupled to the electrical load 523.

Referring to FIG. 17, shown is one example of a phase shifter 516/527.The phase shifter 516/527 may include an active circuit 506 thatsynthesizes an inductance L as can be appreciated. In this respect, theimpedance presented by the phase shifter 516/527 may be adjusted asneeded. Specifically, the capacitors CS and the active circuit 506 maybe adjustable or variable as shown to facilitate changing the impedancepresented by the phase shifter 516/527 as can be appreciated.

Referring next to FIG. 18, shown is an example of cascaded second orderlow pass filter cells (LPF cells), each LPF cell comprising an inductorL and a capacitor C. Referring back to FIG. 8, the power multiplyingnetwork 203 actually comprises a series of cascaded LPF cells in amultiply-connected structure. According to various embodiments, as willbe described in greater detail, each LPF cell in the power multiplyingnetwork 203 may be replaced with an active filter circuit.

Referring then to FIG. 19, shown is a multiply-connected power processor600 according to various embodiments. The multiply-connected powerprocessor 600 comprises a power processing network 603 that comprises aplurality of cascaded active circuits 606. Each of the active circuits606 mimic a passive circuits such as the LPF cells mentioned above.Specifically, the active circuits 606 may each mimic an LPF cell, forexample, or other passive circuit inherent in the power multiplyingnetwork 203 (FIG. 8).

The multiply-connected power processor 600 includes a source coupling613 and a load coupling 616. The source coupling 613 couples an AC powersource 619 to the power processing network 603. The AC power source 619may operate at power frequencies (i.e. 50 Hz., 60 Hz., 400 Hz.) or anyother frequency for which the multiply-connected power processor 600 isdesigned. Similarly, the load coupling 616 couples the electrical load623 to the power processing network 603. The source coupling 613 couplesAC power from the AC power source 619 into the multiply-connected powerprocessor 600 in a manner similar to the couplers described above.Similarly, the load coupling 616 diverts AC power out of the powerprocessing network 603 to the electrical load 623. Associated with thesource coupling 613 and the load coupling 616 is a phase shifter 626that may be employed to adjust the impedance presented by the source andload couplings 613 and 616 as seen by the power processing network 603in order to ensure that the electrical length of the power processingnetwork 603 is a multiple of the wavelength of a power signal at theoperating frequency of the multiply-connected power processor 600. Apower source V_(S) is applied as the active circuit 606. The powersource V_(S) is essentially the same as the power sources V_(S)described above with respect to the multiply-connected power processor500 (FIG. 13).

The multiply-connected power processor 600 further includes a powerprocessor controller 633 that receives various inputs and generatesvarious outputs to control the operation of the multiply-connected powerprocessor 600. In particular, the power processor controller 633receives an input from the power processing network 603 that indicatesthe magnitude of the power stored within the power processing network603 at any given moment. Also, the power processor controller 633receives an input indicating a magnitude of the voltage at the inputport of the source coupling 613 that indicates a state of the AC powergenerated by the AC power source 619. This input informs the powerprocessor controller 633 of the existence of blackouts, brownouts,voltage surges, voltage sags, transient voltages, and other anomaliesexperienced in the AC power received by the multiply-connected powerprocessor 600. The power processor controller 633 further receives aninput indicating a state of the load 623.

During operation of the multiply-connected power processor 600, thepower processor controller 633 may be configured for various differentmodes of operation depending upon the purpose of the multiply-connectedpower processor 600. To this end, the power processor controller 633reacts to one or more of the inputs to control the operation of themultiply-connected power processor 600. For example, the power processorcontroller 633 can control various components of the multiply-connectedpower processor 600 in order to vary the power flowing to the load 623,and/or to vary the amount of power stored in the power processingnetwork 603. The total amount of power stored in the power processingnetwork 603 is limited by the current carrying capacity or power ratingof the active circuits 606 which can otherwise become saturated as canbe appreciated.

In order to control the level of power stored in the power processingnetwork 603, the power processor controller 633 includes outputs tocontrol the operation of the source coupling 613, the amplifier gain ofthe active circuits 606, and the operation of the load coupling 616. Byvirtue of the control signal generated by the power processor controller633 and applied to the source coupling 613, the power processorcontroller 633 controls the rate at which power is coupled into thepower processing network 603. Also, a second control signal generated bythe power processor controller 633 and applied to the source coupling613 controls the rate at which power is coupled into the powerprocessing network 603. The power processor controller 633 generates acontrol signal that is applied to each of the active circuits 606 todynamically control an amplifier gain of the active circuits 606. Whencontrolling the amplifier gain of the active circuits 606, such circuitsare controlled so as to operate in a linear range to avoid saturation ascan be appreciated.

Assuming that the multiply-connected power processor 600 is employed asan uninterruptible power supply, for example, then the power processorcontroller 633 may act to ensure that a minimum amount of power isstored in the power processing network 603 at any given time. Assuming,for example, that the power supply V_(S) comprises DC batteries, thepower processor controller 633 is configured to ensure that a cleanpower signal is applied to the electrical load 623 from the loadcoupling 616. To the extent that the electrical load 623 increases atany given time, then the power processor controller 633 may increase theamount of power coupled into the power processing network 603 by virtueof control of the source coupling 613 to ensure that an adequate amountof power is applied to the electrical load 623 while, at the same time,maintaining a desired power level in the power processing network 603.

In the event that the AC power source 619 is lost, then the powerprocessor controller 633 may increase the amplifier gain of the activecircuits 606, thereby drawing power from the DC batteries or other powersource in order to maintain the desired level of power within the powerprocessing network 603. Of course, if the battery source V_(S) isdepleted and the AC power source 619 remains in a blackout state, thenit is possible that the multiply-connected power processor 600 may runout of power to supply the electrical load 623.

The multiply-connected power processor 600 also acts as a narrow bandpass filter at the power frequency for which it is designed.Specifically, given that the active circuits 606 synthesize a passivelumped element circuits in the power processing network 603, then thepower processing network 603 is a resonant circuit with a resonantfrequency at the frequency of the power signal for which it is designedas described above. As such, the power processing network 603 acts as anarrow band-pass filter at its resonant frequency. Consequently, signalsimposed on a power signal from the AC power source 619 such as highfrequency voltage transients and the like that enter the powerprocessing network 603 through a source coupling 613, for example, aredissipated in the power processing network 603. The energy associatedwith such transients or other anomalies may be lost as heat or may betransformed into the power signal propagating in the power processingnetwork 603.

In addition, short term brownouts or voltage sags that last a few cyclesof the power signal from the AC power source 619 may be compensated forusing power stored in the power processing network 603 at any giventime. Also, the power processor controller 633 may be configured torespond to such anomalies by increasing the amplifier gain of the activecircuit 606 during the duration of such anomalies so as to maintain aconstant power level within the power processing network 603 in spite ofa temporary short term loss of the AC power source 619 or uponexperiencing a voltage sag in such power source.

Thus, by virtue of the power storage characteristics, the resonantnature of the power processing network 603, and the ability to vary theamplifier gain of the active circuit 606, the multiply-connected powerprocessor 600 may provide clean power to the electrical load 623 withoutinterruption even when the AC power source 619 experiences brownouts,voltage sags, voltage surges, transient voltages, and other problems.

In addition, it may be the case that voltage surges occur in the ACpower applied to the source coupling 613. The power processor controller633 may be configured to maintain the power level in the powerprocessing network 603 under its the maximum power storage capacity.This would allow for room to absorb voltage surges experienced in the ACpower source into the power processing network 603. Also, the sourcecoupling 613 may be controlled to reduce the rate that power enters thepower processing network 603 upon detection of a voltage surge to ensurethat the power level within the power processing network 603 does notexceed the maximum capacity of the power processing network 603. Also,the load coupling 616 may be controlled to ensure that the powerdirected to the electrical load 623 substantially equals the powerdemand of the electrical load 623 at any given instant.

Still further, the multiply-connected power processor 600 may beemployed in another application as a node to supply the input of powerof a structure such as a building, residence, or other structure. Insuch an embodiment, the power source V_(S) may be supplied by arenewable source such as, for example, solar panels, windmills, or othersource. In such case, it may be desirable to maximize the amount ofpower from such renewable sources. To this end, the power processorcontroller 633 may increase the amplifier gain of the active circuits606 to maximize the amount of power generated in the power processingnetwork 603 from the renewable source, thereby minimizing the amount ofpower coupled into the power processing network 603 from the AC powersource 619.

In this respect, the multiply-connected power processor 600 provides adegree of flexibility in that the AC power source 619 may be drawn upononly as needed in situations where such renewable sources may besomewhat unreliable. For example, cloud cover may inhibit theeffectiveness of solar cells from time to time, or windmills may sitstill due to a lack of wind. In this respect, the multiply-connectedpower processor 600 may be employed to reduce reliance upon fossil fuelssuch as coal burned in coal fired generating stations and the like ascan be appreciated. Further, if high efficiency operational amplifiersare employed within the active circuit 606, it is possible that agreater amount of power may be captured from the renewable voltagesource V_(S) than would be captured by traditional DC to AC conversioncircuits.

In addition, several AC power sources may be coupled to themultiply-connected power processor 600. Such AC power sources mayinclude the AC power generation source 619 or other sources. Forexample, other source couplings 613 may be employed to couple other ACpower sources to the power processing network 603. Such other AC powergeneration sources may comprise, for example, windmills, portablegenerators, or other types of power generation sources as can beappreciated. The power processing network 603 provides for a degree ofisolation between the AC power source 619 and other power sourcescoupled thereto through other source couplings 613. In this respect,once a power signal enters the power processing network 603, it iseffectively decoupled from the power generating source 619 thatoriginally generated such signal. As such, the power processing network603 offers a degree of electrical isolation between power sourcescoupled to the power processing network 603 through source couplings613.

For example, given that the power signal in the power processing network603 is independent of the AC power signal generated by the AC powersource 609, then any other power generating source coupled to the powerprocessing network 603 by a source coupling 613 need only besynchronized with the power signal that travels around the powerprocessing network 603. That is to say, once a power signal propagatesthrough the power processing network 603, it propagates through thepower processing network 603 independent of the power generating source.As such, the power signal can provide an independent reference forfrequency synchronization for any other power source coupled to thepower processing network 603. Stated another way, the power signalpropagating around the power processing network 603 becomes a frequencyreference for power sources coupled thereto rather than the signalgenerated by any one power source.

This ultimately will minimize any negative effect on frequencysynchronization that a secondary power source may have on the operationof a AC power generation source 619. This reflects the fact that for agiven power grid, secondary sources of power generation cannot make upmore than 20 to 25 percent of the total generation on the grid withoutnegatively affecting the ability to maintain frequency synchronization.However, the multiply-connected power processor 600 allows secondarysources to be coupled thereto to supply power to the load 623 withoutnegatively affecting potential frequency synchronization on a power gridcoupled to the multiply-connected power processor 600.

Rather, such secondary generation sources coupled to the powerprocessing network 603 need only be synchronized with the signal as ittravels around the power processing network 603 independent of thesignal generated by the AC power generation source 609 that has not yetentered the power processing network 603.

Further, the impedance synthesized by the active circuits 606 may bevaried parametrically by varying components such as capacitancesassociated with such active circuits as can be appreciated.

With reference to FIG. 20, shown is one example of an active circuit 606according to various embodiments. The active circuit 606 depicted inFIG. 20 comprises a Sallen-Key active filter. The Sallen-Key filter isbut one of many different types of circuit configurations that may beemployed as active circuits 606 according to the various embodiments.For example, other common circuit configurations that may be employed asthe active circuits 606 include, for example, Butterworth, Bessel,Chebyshev, State-Variable, Generalized Impedance Converter (GIC)filters, and many other circuit configurations.

With reference to FIG. 21, shown is one example of the source coupling613. As shown, the source coupling includes coupling delay activecircuits 653 and 656. The coupling delay active circuits 653 and 656provide a quarter wavelength delay for the power signal generated by theAC power source 619 as described above. Also, the coupling capacitorsC_(PX) and C_(PY) are variable capacitors that are controlled by thepower processor controller 633 to control a rate at which power entersthe power processing network 603 from the AC power source 619.

With reference to FIG. 22, shown is an example of the load coupling 616that is controlled by the power processor controller 633 to determinethe rate at which power is diverted from the power processing network603 to the electrical load 623 (FIG. 19). The load coupling 616 includesthe coupling delay active circuits 663 and 666 that provide the quarterwavelength delay of the power signal in order to facilitate thedirectional coupler capability of the load coupling 616. Similarly, theload coupling 616 includes the coupling capacitors C_(L1) and C_(L2)that are controlled by the power processor controller 633 to determinethe rate at which power exits the power processing network 603 and isdiverted to the load 623.

With reference to FIGS. 21 and 22, the coupling delay active circuits653, 656, 663, and 666 each comprise active circuits such as the activecircuits 606 as described above and synthesize passive impedances asdescribed above. Thus, the size of the source coupling 613 and the loadcoupling 616 are reduced accordingly.

With reference to FIG. 23, shown is a multiply-connected power processor700 according to various embodiments that is employed for the conversionof DC power to AC power, or for the conversion of an AC power signalfrom one frequency to another.

The multiply-connected power processor 700 includes essentially the samecomponents of the multiply-connected power processor 600 described abovewith reference to FIG. 19. However, the multiply-connected powerprocessor 700 further includes a signal generator 703 that is coupled tothe power processing network 603 via the source coupling 613.

The signal generator 703 is configured to generate a signal at a desiredfrequency such as a power frequency or any other frequency for which themultiply-connected power processor 700 is designed. The signal generatedby the signal generator 703 is employed to “seed” or prime a powersignal in the power processing network 603. The signal generated by thesignal generator 703 is coupled into the power processing network 603 bythe source coupling 703. As the signal propagates around the powerprocessing network 603, the amplifier gain of the active circuits 606 iscontrolled by the power processor controller 633 to amplify the signalso that a desired power level in the power processing network 603 isachieved. Once the power signal is established around the powerprocessing network 603, then the signal generator 703 may be turned off.

When power is diverted to the electrical load 623 via the load coupling616, the power processor controller 633 responds by increasing theamplifier gain to maintain the power level within the power processingnetwork 603 at a desired constant level. Where the power source V_(S)comprises a DC power source, the multiply-connected power processor 700may be employed as an AC to DC converter without typical AC to DCconversion circuitry. Where the power source V_(S) comprises an ACsource, the multiply-connected power processor 700 can be employed forfrequency conversion where the frequency of the power source V_(S)differs from the resonant frequency of the multiply-connected powerprocessor 700. To increase the efficiency of the multiply-connectedpower processor 700, high efficiency operational amplifiers may beemployed.

The power processor controller 706 is configured to monitor theelectrical load 623 and the level of power in the power processingnetwork 603 and control the amplifier gain of the active circuits 606and the load coupling 616 to generate and divert the proper level ofpower to the electrical load 623. The power processor circuit 706further controls the signal generator 703 to generate a signal to seedor prime a signal in the power processing network 603 during startup asdescribed above.

It should be emphasized that the above-described embodiments of thepresent invention are merely possible examples of implementations,merely set forth for a clear understanding of the principles of theinvention. Many variations and modifications may be made to theabove-described embodiment(s) of the invention without departingsubstantially from the spirit and principles of the invention. All suchmodifications and variations are intended to be included herein withinthe scope of this disclosure and the present invention and protected bythe following claims.

Therefore, having thus described the invention, at least the followingis claimed:
 1. A method, comprising: receiving, by a control systemcoupled to a load coupler, an input signal associated with an electricalload coupled to a multiply-connected velocity inhibiting circuit via theload coupler, the input signal indicating a magnitude of the electricalload; and determining, by the control system, a determined rate at whichalternating current (AC) power is diverted from the multiply-connectedvelocity inhibiting circuit to the electrical load based at least inpart upon the input signal.
 2. The method of claim 1, further comprisingadjusting a rate at which AC power is diverted to the electrical load bythe load coupler based at least in part upon the determined rate.
 3. Themethod of claim 2, wherein the rate at which the AC power is diverted tothe electrical load is adjusted by the load coupler in response to acontrol signal generated by the control system.
 4. The method of claim1, further comprising adjusting a rate at which AC power enters themultiply-connected velocity inhibiting circuit from an AC power sourcevia a source coupler in response to the determined rate.
 5. The methodof claim 4, wherein the rate at which the AC power enters themultiply-connected velocity inhibiting circuit from the AC power sourceis based at least in part upon the determined rate and an amount ofpower stored in the multiply-connected velocity inhibiting circuit. 6.The method of claim 5, wherein the rate at which the AC power enters themultiply-connected velocity inhibiting circuit from the AC power sourceis adjusted to maintain the amount of power stored in themultiply-connected velocity inhibiting circuit at a substantiallyconstant level.
 7. The method of claim 5, wherein the rate at which theAC power enters the multiply-connected velocity inhibiting circuit fromthe AC power source is adjusted to maintain the amount of power storedin the multiply-connected velocity inhibiting circuit at or above apredefined minimum amount of power.
 8. The method of claim 1, furthercomprising synthesizing at least one passive lumped element in themultiply-connected velocity inhibiting circuit using an active circuit.9. The method of claim 8, further comprising dynamically controlling again of the active circuit.
 10. The method of claim 7, furthercomprising generating a negative resistance in the multiply-connectedvelocity inhibiting circuit by synthesizing a parametric reactance asthe at least on passive lumped element.
 11. A method, comprising:receiving, by a control system coupled to a source coupler, an inputsignal associated with an alternating current (AC) power source coupledto a multiply-connected velocity inhibiting circuit via the sourcecoupler, the input signal indicating a state of AC power provided fromthe AC power source; and determining, by the control system, adetermined rate at which the AC power enters the multiply-connectedvelocity inhibiting circuit from the AC power source based at least inpart upon the input signal.
 12. The method of claim 11, furthercomprising adjusting a rate at which the AC power enters themultiply-connected velocity inhibiting circuit via the source couplerbased at least in part upon the determined rate.
 13. The method of claim12, wherein the rate at which the AC power enters the multiply-connectedvelocity inhibiting circuit is adjusted by the source coupler inresponse to a control signal generated by the control system.
 14. Themethod of claim 12, wherein the rate at which the AC power enters themultiply-connected velocity inhibiting circuit from the AC power sourceis based at least in part upon the determined rate and an amount ofpower stored in the multiply-connected velocity inhibiting circuit. 15.The method of claim 14, wherein the rate at which the AC power entersthe multiply-connected velocity inhibiting circuit from the AC powersource is adjusted to prevent the amount of power stored in themultiply-connected velocity inhibiting circuit from exceeding apredefined storage limit of the multiply-connected velocity inhibitingcircuit.
 16. The method of claim 14, wherein the rate at which the ACpower enters the multiply-connected velocity inhibiting circuit from theAC power source is adjusted to maintain the amount of power stored inthe multiply-connected velocity inhibiting circuit at a substantiallyconstant level.
 17. The method of claim 11, wherein the input signalcomprises a voltage magnitude at an input of the source coupler.
 18. Themethod of claim 11, further comprising adjusting a rate at which ACpower is diverted from the multiply-connected velocity inhibitingcircuit to an electrical load by a load coupler in response to thedetermined rate.
 19. The method of claim 18, wherein the rate at whichthe AC power is diverted from the multiply-connected velocity inhibitingcircuit from the multiply-connected velocity inhibiting circuit is basedat least in part upon the determined rate and an amount of power storedin the multiply-connected velocity inhibiting circuit.
 20. The method ofclaim 11, further comprising synthesizing at least one passive lumpedelement in the multiply-connected velocity inhibiting circuit using atleast one active circuit.